Open Access

Unified commutation-pruning technique for efficient computation of composite DFTs

  • David E. Castro-Palazuelos1, 2Email author,
  • Modesto Gpe. Medina-Melendrez1,
  • Deni L. Torres-Roman2 and
  • Yuriy V. Shkvarko2
EURASIP Journal on Advances in Signal Processing20152015:100

https://doi.org/10.1186/s13634-015-0285-z

Received: 30 March 2015

Accepted: 11 November 2015

Published: 26 November 2015

Abstract

An efficient computation of a composite length discrete Fourier transform (DFT), as well as a fast Fourier transform (FFT) of both time and space data sequences in uncertain (non-sparse or sparse) computational scenarios, requires specific processing algorithms. Traditional algorithms typically employ some pruning methods without any commutations, which prevents them from attaining the potential computational efficiency. In this paper, we propose an alternative unified approach with automatic commutations between three computational modalities aimed at efficient computations of the pruned DFTs adapted for variable composite lengths of the non-sparse input-output data. The first modality is an implementation of the direct computation of a composite length DFT, the second one employs the second-order recursive filtering method, and the third one performs the new pruned decomposed transform. The pruned decomposed transform algorithm performs the decimation in time or space (DIT) data acquisition domain and, then, decimation in frequency (DIF). The unified combination of these three algorithms is addressed as the DFTCOMM technique. Based on the treatment of the combinational-type hypotheses testing optimization problem of preferable allocations between all feasible commuting-pruning modalities, we have found the global optimal solution to the pruning problem that always requires a fewer or, at most, the same number of arithmetic operations than other feasible modalities. The DFTCOMM method outperforms the existing competing pruning techniques in the sense of attainable savings in the number of required arithmetic operations. It requires fewer or at most the same number of arithmetic operations for its execution than any other of the competing pruning methods reported in the literature. Finally, we provide the comparison of the DFTCOMM with the recently developed sparse fast Fourier transform (SFFT) algorithmic family. We feature that, in the sensing scenarios with sparse/non-sparse data Fourier spectrum, the DFTCOMM technique manifests robustness against such model uncertainties in the sense of insensitivity for sparsity/non-sparsity restrictions and the variability of the operating parameters.

Keywords

Composite length discrete Fourier transformDecimationDecompositionFast Fourier transformPruning

1 Introduction

1.1 Motivation

Many signal processing applications require computation of the so-called pruned discrete Fourier transform (DFT), i.e., an efficient alternative to compute the required DFT when the input sequence and/or the required output sequences are smaller than the length of the full DFT (a full DFT means that all the output components are to be computed, and all the input elements are used to compute the transform); in the literature those are referred to as pruned fast Fourier transforms (FFTs) or pruned DFTs [1]. Common practical examples relate to, e.g., the least mean squared (LMS) optimal DFT-based pruned signal filtering [2], and the complexity-reduced computational implementation of the orthogonal frequency division multiplexing systems [3]. Another practical example relates to efficient implementation of the matched spatial filtering (MSF) algorithm for performing the range and azimuth data compression in unfocused of fractionally focused synthetic aperture radar (SAR) system that both employ the pruned DFT-based MSF processing of the trajectory data signals performed in a factorized fashion in the so-called slow time and fast time data acquisition scales [46]. Other examples relate to DFT-based analysis of remote sensing (RS) data acquired with a variety of sensor systems, ranging from seismology [7] to multispectral radiometry [8]. Other authors as Zhu et al. in [9] proposed an algorithm for performing SAR polar format re-gridding interpolation suited for the logic-in-memory paradigm (hardware/architecture solution) and to provide the necessary design automation tool chain to implement their proposed algorithm (e.g., FFTs for image formation) in advanced silicon technology. It is important to note that a majority of real-world RS data acquisition and processing problems can be qualified as sensing in harsh environments [48, 10, 11] in the sense of intrinsic problem model uncertainties peculiar for such RS modalities. In a context of pruned DFTs, realistic harsh sensing scenarios are characterized by the uncertainties attributed to zero-padded input data acquisition modes with variable composite length windowing of the input and/or output Fourier transform sequences, in general cases, with non-sparse Fourier spectra [1012]. Those specifics motivate the development of efficient pruned DFT/FFT techniques particularly adapted for computational implementation with uncertain data acquired in harsh sensing scenarios.

1.2 Related work

Traditional DFT algorithms adapted for such uncertain scenarios typically employ some pruning methods without any commutations, which prevent them from attaining the potential computational efficiency. Most of the proposals reported in the literature are based on construction of pruning modalities of specific FFT-related algorithms. Some of them prune the input of a specific FFT algorithm, others prune the output, and just a few can prune the input and output (input-output) at the same time. Markel in [1], and Skinner in [13], proposed the input pruning methods based on a radix-2 FFTs, while Yuan et al., in [14], proposed an input pruning of a split-radix FFT. The approaches of Bouguezel et al. [15] and Fan et al. [16] are applicable for output pruning a radix-2 FFTs, while the Xu’s et al. [3] proposal suggests pruning the output of a split-radix FFT. In addition, Sreenivas et al., in [17], Roche, in [18], and Wang et al., in [19], developed the methods for pruning the input-output at the same time. The first one is based on a radix-2 FFT, the second one employs the split-radix FFT, and the third one performs the mixed-radix FFT, respectively. A majority of those methods are applicable only for computing DFTs with the length of a power of two that drastically restricts their applicability to general uncertain sensing scenarios.

On the other hand, a family of novel so-called sparse FFT (SFFT) algorithms adapted to computing the FFTs, when only a few Fourier spectrum coefficients of the input signal are different from zero (few largest coefficients of the Fourier transform spectrum), has been developed recently [20, 21]. The celebrated SFFT-related algorithms, so-called SFFTv1 and SFFTv2, were reported by Hassanieh et al., in [20]. Later, in [21], the improved SFFT-related versions, addressed as SFFTv3 and SFFTv4, were reported. Another algorithm that considers the Fourier spectrum sparsity restrictions is the so-called FADFT-2 reported and implemented in the AAFFT library [22]. However, the SFFT-related algorithms significantly outperform the AAFFT as it was corroborated in [20].

It is worthwhile to mention that the SFFT-related techniques are applicable only for the sparse sensing scenarios; e.g., referring to [20, 21], the authors exemplified the sparsity level by imposing the restriction that up to 89 % of the Fourier coefficients are zeroes or negligible, thus can be discarded. Such a restriction could be valid in a variety of data compressing applications, e.g., compression and recovery of video data not degraded by noise and/or imaging system instrumental function [20]. Nevertheless, the restriction on such sparsity is not valid for many real-world operational scenarios, e.g., processing of the RS data acquired in harsh sensing environments [48, 1012]. For example, in SAR imaging of non-homogeneous scenes, e.g., urban areas, non-uniformly textured zones, etc., a majority of the Fourier transform coefficients should be considered for feature-enhanced MSF-based imaging [5, 6]; thus, an 89 % of sparsity level restriction is never a feasible model assumption.

In this paper, we are interested in developing the pruned DFT (DFTs of highly composite length) algorithms applicable for near-real-time signal processing and analysis in uncertain sensing scenarios (i.e., with non-guaranteed sparsity of the data Fourier spectra); that is why the family of the SFFT-related techniques is beyond our detailed study here. Nevertheless, for the purpose of generality, in Section 4, we perform comparative analysis of our developed methods with the SFFT under the same conditions and constraints for different combinations of the specified processing/operational parameters.

In the related literature, in which the feasible non-sparse scenarios are considered, two competing approaches for pruning the composite (no prime) length DFTs were addressed. Sorensen et al., in [23], proposed two methods to prune composite length DFTs, first one to prune the input and another one to prune the output. Next, the methodology of Medina-Melendrez et al., in [24], merges the methods developed originally in [23] to obtain a composite structure that is capable to prune the input and/or output of a general decomposed transform at the same time. It was demonstrated, in [24], that such a computational structure could be as efficient as the one based on specific FFT algorithms [1517]. In [24], a new methodology for decomposition over a composite length DFT has been proposed as a modification of the Sorensen’s approach [23]. Furthermore, the [24] suggests, first, to perform decimation in frequency (DIF) and, second, a decimation in time (DIT). For processing of spatial data, the corresponding decimation in the space domain should be performed similarly to the DIT operation for time data processing. To avoid misunderstandings, in the rest of the paper, we will use the same abbreviation (DIT) for both processing models and consider the time data processing as a principal model. Nevertheless, all developments are directly transferable for the space data processing scenario.

Hence, the three basic stages to compute the composite length DFTs of non-sparse data encompass the input, the intermediate, and the output stages. The decomposed transform is then pruned by eliminating, from the input and output stages, additions and multiplications by zero, multiplications by one, and all other computations not needed to obtain the required Fourier transform coefficients. In [24], such the multistage decomposed and pruned transform is referred to as FFTDIF−DIT−TD (here, that method is referred as DFTDIF−DIT−Pr). Nevertheless, both methods addressed in [23, 24] do not achieve the lowest attainable number of the required arithmetic operations. A possible alternative for computing few Fourier coefficients from few input elements (all non-zero, thus non-sparse) can be addressed based on the application of the second-order Goertzel algorithm [23] modified to accept the input elements in a reverse order.

1.3 Novel contributions

The main contribution of this paper consists in the development of a new alternative method for efficient computing of a composite length DFT, when the input sequence and/or the required output sequence are smaller than the length of the full transform. Our proposal guarantees the same or smaller number of arithmetic operations in comparison with the competing methods in the literature. Moreover, it manifests robustness against sparsity/non-sparsity restrictions and the variability of the operating parameters as detailed in Sections 3 and 4.

The innovative idea is to automatically commute among three modalities to implement the DFT: the direct method, the recursive method, and the pruned decomposed transform. Thus, our new proposed composite approach unifies the decomposition of the DFT with its pruning. First, we develop an alternative technique to compute the pruned decomposed transform, in which the DIT is performed at the first stage followed by the DIF. We address this method as DFTDIT−DIF−Pr. An analysis of the two alternatives (DFTDIT−DIF−Pr and DFTDIF−DIT−Pr) verifies that the DFTDIT−DIF−Pr requires a smaller or as maximum equal number of arithmetic operations compared with the DFTDIF−DIT−Pr, so the use of the DFTDIT−DIF−Pr is strongly recommended when the decomposed and pruned transforms are required. Next, we demonstrate that our proposal requires a lower number of arithmetic operations than any of the pruning-based competing methods [3, 14, 23, 24]. Further, we demonstrate that both decomposed transforms (DFTDIF−DIT−Pr and DFTDIT−DIF−Pr) can be obtained from a general decomposition methodology. Also, it manifests the robustness in sparse and non-sparse sensing scenarios (i.e., operability for an arbitrary number of consecutive input elements (L i ), the number of consecutive outputs that should be computed (L o ), and the length of the full transform (N)) in contrast to the recently developed most prominent SFFT family-related methods [20, 21] operable in sparse scenarios only.

It is noteworthy to mention that in the majority of practical computational scenarios, significant savings in the number of arithmetic operations with the proposed technique are achieved, e.g., in Section 4.1, the DFTCOMM technique compared with split-radix FFT (SRFFT) algorithm produces savings of 42 to 92 %.

The rest of the paper is organized as follows: in Section 2, the general decomposition transform methodology is described and explained. An analysis of all feasible transform decomposition methods is presented next in Section 3 followed by the combinational hypotheses testing optimization-based selection of the best decomposition transform permutation modality that yields the unified commutation-pruning DFTCOMM technique. In Section 4, comparisons among the developed unified commutation-pruning technique and other competing algorithms in the sense of savings in the number of required arithmetic operations are presented and featured. Also, the proposed DFTCOMM method is compared in detail with the most prominent competing SFFT-related algorithms in the context of computing the DFTs in both sparse and non-sparse (harsh) sensing scenarios for different values of the operational parameters (L i , L o , and N). Concluding remarks in Section 5 summarize the study. The Appendix provides a pseudo-code for implementing the proposed method.

2 DFT transform decomposition

The definition of the DFT of a sequence of length N (DFT N ) is given by
$$ X(k)={\displaystyle \sum_{n=0}^{N-1}x(n){W}_N^{nk}}\kern0.5em \mathrm{f}\mathrm{o}\mathrm{r}\kern0.75em k=0,1,2,\dots, N-1 $$
(1)
where \( {W}_N^{nk}={e}^{-j2\pi nk/N} \) is the kernel of the transform. Let us define L i as the number of consecutive input elements different from zero and L o as the number of consecutive outputs that should be computed. If N is a composite number formed by multiplications of many integer factors, the DFT N can be decomposed into smaller DFTs. In particular, the DFT N can be decomposed into three stages of DFTs (an input stage, an intermediate stage, and an output stage) in order to avoid the arithmetic operations involving zeros, multiplications by one, and the operations not required to compute the final outputs. Here beneath, we briefly describe such feasible decompositions. Assuming that there are two integer factors, D ip and D op , of N such that N/D ip D op  P is an integer, the indexes n and k can be re-expressed as
$$ n={n}_1+{D}_{op}{n}_2+\left(\frac{N}{D_{ip}}\right){n}_3\kern1em \mathrm{f}\mathrm{o}\mathrm{r}\kern0.75em \left\{\begin{array}{l}{n}_1=0,1,2,\dots, {D}_{op}-1\\ {}{n}_2=0,1,2,\dots, {\scriptscriptstyle \raisebox{1ex}{$N$}\!\left/ \!\raisebox{-1ex}{${D}_{ip}{D}_{\mathrm{op}}$}\right.}-1\\ {}{n}_3=0,1,2,\dots, {D}_{ip}-1\end{array}\right. $$
(2)
$$ k={k}_1+{D}_{ip}{k}_2+\left(\frac{N}{D_{op}}\right){k}_3\kern1em \mathrm{f}\mathrm{o}\mathrm{r}\kern1em \left\{\begin{array}{l}{k}_1=0,1,2,\dots, {D}_{ip}-1\\ {}{k}_2=0,1,2,\dots, {\scriptscriptstyle \raisebox{1ex}{$N$}\!\left/ \!\raisebox{-1ex}{${D}_{ip}{D}_{op}$}\right.}-1\\ {}{k}_3=0,1,2,\dots, {D}_{op}-1.\end{array}\right. $$
(3)
Substituting n and k in (1) by (2), (3), the original DFT N is decomposed into
$$ \begin{array}{l}X\left({k}_1+{D}_{ip}{k}_2+\frac{N}{D_{op}}{k}_3\right)={\displaystyle \sum_{n_1=0}^{D_{op}-1}{\displaystyle \sum_{n_2=0}^{P-1}{\displaystyle \sum_{n_3=0}^{D_{ip}-1}x\left({n}_1+{D}_{op}{n}_2+\frac{N}{D_{ip}}{n}_3\right)}}}\\ {}\kern10.5em \times {W}_N^{\left({n}_1+{D}_{op}{n}_2+\left(N/{D}_{ip}\right){n}_3\right)\left({k}_1+{D}_{ip}{k}_2+\left(N/{D}_{op}\right){k}_3\right)}.\end{array} $$
(4)
Here, it is assumed that D ip and D op are chosen in such a way that N/D ip  ≥ L i and N/D op  ≈ L o . Thus, index n 3 is always equal to zero; k 3 is near 0, hence (4) can be next rewritten as follows
$$ \begin{array}{l}X\left({k}_1+{D}_{ip}{k}_2+\frac{N}{D_{op}}{k}_3\right)={\displaystyle \sum_{n_1=0}^{D_{op}-1}{\displaystyle \sum_{n_2=0}^{P-1}x\left({n}_1+{D}_{op}{n}_2\right){W}_N^{\left({n}_1+{D}_{op}{n}_2\right)\left({k}_1+{D}_{ip}{k}_2+\left(N/{D}_{op}\right){k}_3\right)}}}\\ {}={\displaystyle \sum_{n_1=0}^{D_{op}-1}{\displaystyle \sum_{n_2=0}^{P-1}x\left({n}_1+{D}_{op}{n}_2\right){W}_N^{\left({n}_1{k}_1+{D}_{op}{n}_2{k}_1+{D}_{ip}{n}_1{k}_2+{D}_{ip}{D}_{op}{n}_2{k}_2+{n}_1{k}_3N/{D}_{op}+{n}_2{k}_3N\right)}}}.\end{array} $$
(5)
The computation of (5) is more efficient than the direct computation of the DFT N since the complex arithmetic operations dependent on n 3 have been pruned. The complex exponential in (5) can next be grouped in different ways, resulting in different structures for the pruned decomposed transform. The methodology of [24] suggests expressing the pruned decomposed transform as
$$ \begin{array}{l}X\left({k}_1+{D}_{ip}{k}_2+\frac{N}{D_{op}}{k}_3\right)={\displaystyle \sum_{n_1=0}^{D_{op}-1}\left\{{\displaystyle \sum_{n_2=0}^{P-1}\left[{W}_N^{\left(\left({n}_1+{D}_{op}{n}_2\right){k}_1\right)}x\left({n}_1+{D}_{op}{n}_2\right)\right.}\right.}\\ {}\kern10.5em \times \left.\left.{W}_{N/{D}_{ip}{D}_{op}}^{n_2{k}_2}\right]\right\}{W}_N^{\left({n}_1\left({D}_{ip}{k}_2+{k}_3N/{D}_{op}\right)\right)}.\end{array} $$
(6)
The pruned decomposed transform of (6) can be interpreted as follows: first, apply DIF to the DFT N with D ip as a decomposition factor, then, DIT to the resulting DFTs with D op as a decomposition factor and, finally, perform the pruning. In [24], the pruned decomposed transform of (6) was addressed as an FFTDIF−DIT−TD modality, that in our notations, we refer to as DFTDIF−DIT−Pr. A computational diagram of such technique (6) is presented in Fig. 1.
Fig. 1

General computational diagram of the DFTDIF−DIT−Pr (referred as FFTDIF−DIT−TD in [24, Fig. 1])

An alternative grouping of the complex exponentials in (5) yields
$$ \begin{array}{l}X\left({k}_1+{D}_{ip}{k}_2+\frac{N}{D_{op}}{k}_3\right)={\displaystyle \sum_{n_1=0}^{D_{op}-1}\left\{{\displaystyle \sum_{n_2=0}^{P-1}\left[{W}_N^{\left({D}_{op}{n}_2{k}_1\right)}x\left({n}_1+{D}_{op}{n}_2\right){W}_P^{n_2{k}_2}\right]}\right\}}\\ {}\kern10.25em \times {W}_N^{\left({n}_1\left({k}_1+{D}_{ip}{k}_2+{k}_3N/{D}_{op}\right)\right)}\\ {}\kern9em ={\displaystyle \sum_{n_1=0}^{D_{op}-1}\left\{{\displaystyle \sum_{n_2=0}^{P-1}\left[y\left({n}_1,{n}_2,{k}_1\right){W}_P^{n_2{k}_2}\right]}\right\}}\\ {}\kern10.25em \times {W}_N^{\left({n}_1\left({k}_1+{D}_{ip}{k}_2+{k}_3N/{D}_{op}\right)\right)}.\end{array} $$
(7)

The computing of the pruned decomposed transform (7) requires, first, application of DIT to the DFT N with D op as a decomposition factor and, then, application of DIF to the resulting DFTs with D ip as a decomposition factor.

Hence, we refer to the pruned decomposed transform of (7) as a DFTDIT−DIF−Pr modality. A computational diagram of such the technique (7) is presented in Fig. 2.
Fig. 2

General computational diagram of the DFTDIT-DIF-Pr

The DFTDIT−DIF−Pr involves three processing stages: an input stage (computation of y(n 1, n 2, k 1)), an intermediate stage (computation of D ip D op DFTs of length P), and an output stage (computation of the complex multiplications and additions dependent on index n 1).

3 Proposed method

Our method employs three different alternatives to compute the DFT N : a direct method, a recursive method, and/or a pruned decomposed transform. Admissible permutations/allocations of all feasible decomposition-pruning modalities compose all possible hypotheses regarding the feasible alternative schemes for computing the composite DFTs.

All feasible commuting-pruning implementation structures are listed in Table 1. Those could be addressed as possible search “hypotheses” to be tested. Thus, the problem of selection of an optimal computing-pruning implementation structure can be recast as a hypotheses testing task. All feasible hypotheses relate to formal implementation structures specified in Table 1. Four of them prescribe cascade computational implementation involving cascade combinations of structures (hypotheses H4,…, H7), while four others (hypotheses H9,…, H12) prescribe combinational unions of the previous hypotheses. It is important to remark that (1), (6), and (7) are the mathematical definitions of H8, H4, and H5, respectively. Hence, the decision-making process that is a selection from those feasible operational prescriptions cannot be formalized as an optimization strategy for minimization of some cost function subject to relevant restrictions/constraints specified in a closed analytical form. Thus, due to the composite combinations (hypotheses over hypotheses with cascade interlaces, as in the cases of hypotheses H9,…, H12), the proper selection of the preferable implementation structure cannot be cast as an analytically tractable closed-form optimization problem. Hence, it should be treated as a test of combinations of hypotheses (hypotheses over hypotheses, as in the case of H9,…, H12), sometimes referred to as a combinational (or combinatorial-type) hypothesis testing problem [23, 24]. The global optimal solution to such a kind of problems presumes test of all feasible hypotheses in the list, making the decision in favor of the best one (in the prescribed quality measure), and rejection of all other competing hypotheses [23, 24]. In our particular case, only 12 hypotheses are admissible/feasible; thus the (global) optimal selection of the best possible implementation structure can be found simply via employing the so-called brute force search over complete hypotheses list specified in Table 1.
Table 1

Complete list of hypotheses \( {\left\{{\mathrm{H}}_h\right\}}_{h=1}^{12} \) regarding feasible commuting-pruning implementation structures

Hypotheses

Specifications

H1 : y = Ax

x = input

H2 : y = Bx

y = output

H3 : y = Fx

A = DIF

H4 : y = ABx

B = DIT

H5 : y = BAx

F = 2BF filtering

H6 : y = ABFx

D = DFT N

H7 : y = BAFx

 

H8 : y = Dx

 

H9 : H4 H6

 

H10 : H5 H7

 

H11 : H9 H3 H8

 

H12 : H10 H3 H8

 
Sorensen et al. [23] sketched how to prune the input and output of DFTs using independent allocations listed in Table 1 as H1, H2, and H3 and featured in Fig. 3a. However, the authors of [23] concluded that their pruning method is less efficient than other pruning methods in the cases when both the number of input and output elements are bounded. They recommended turning to the method proposed by Sreenivas et al., in [17], i.e., to prune the input and output of a power of two length FFTs. Furthermore, an efficient input-output pruning method for a power of two length FFTs was proposed by Roche in [18].
Fig. 3

Possible feasible alternative schemes for efficient computation of composite DFTs when the input and/or output sequences are smaller than the length of the full transform. a State of the art. b New feasible implementation structures. Specifications of the blocks labeled as A, B, F, and D are listed in Table 1

Later, a more efficient input-output pruning method for composite length DFTs was developed in [24]. Such commuting between H4 H6 leads to hypothesis H9 as featured in Fig. 3a. In [24], such a technique was constructed as a modification of the transform decomposition proposed originally by Sorensen et al., in [23], but with extra capability to perform the input-output pruning at the same time. Additionally, the computation of each final output employs a commutation between a direct method and the 2BF filtering algorithm, i.e., the 2BF-filtering algorithm is an efficient method for computing a subset of final outputs from their decomposition transform [23, 24].

In our study, two additional feasible hypotheses are devised to perform unified commutation-pruning techniques for efficient computations of composite length DFTs (hypotheses H11 and H12) as reported in Fig. 3b. Therefore, our proposal relates to an adaptive commuting between feasible implementation structures specified by the union of hypotheses H10 H3 H8 that is included in Table 1 as an alternative composite hypothesis H12. A comparison of computational complexities related to implementation of the competing computational structures formalized by hypotheses H9 and H10 (in the number of required arithmetical operations) is reported in Table 2. Also, the relevant comparisons between two other feasible structures specified by hypotheses H11 and H12(referred here as DFTCOMM−DIF−DIT−Pr and DFTCOMM−DIT−DIF−Pr, respectively), are reported in Tables 2 and 3 and Figs. 5af (in the sense of the number of required arithmetic operations).
Table 2

Total number of arithmetic operations required to compute the input and output stages of DFTDIF−DIT−Pr and DFTDIT−DIF−Pr

Pruned decomposed transform

Corresponding term in (8)

Number of required real arithmetic operations

Limiting constraints

DFTDIF−DIT−Pr

OPERinput

6 (L i  − 1) (D ip  − 1)

 

OPERoutput

2L o (D op  − 1)

(L o  ≤ D ip )

  

2L o (D op  − 1) + 6 (L o  − D ip ) (D op  − 1)

(L o  > D ip ) & (D op  < 4)

  

(L o  − D ip ) (2D op  + 2) + 2D ip (D op  − 1) + (L o  − D ip ) (4D op  − 2) a

(L o  > D ip ) & (D op  ≥ 4)

DFTDIT−DIF−Pr

OPERinput

0

(L i  ≤ D op )

  

6 (L i  − D op ) (D ip  − 1)

(L i  > D op )

 

OPERoutput

6 (L o  − 1) (D op  − 1) + 2L o (D op  − 1)

(D op  < 4)

  

(L o  − 1) (2D op  + 2) + 2 (D op  − 1) + (L o  − 1) (4D op  − 2) a

(D op  ≥ 4)

a Number of required operations when the 2BF filtering method is employed [23, 24]

Table 3

Total number of arithmetic operations required to compute the input and output stages of DFTCOMM−DIF−DIT−Pr and DFTCOMM−DIT−DIF−Pr modalities

Method to compute the DFT N

Number of arithmetic operations

Limiting conditions

Direct method

6 (L o  − 1) (L i  − 1) + 2L o (L i  − 1)

((L i  ≤ D op )|(L o  ≤ D ip )) & (L i  < 4)

2BF filtering method

(L o  − 1) (2L i  + 2) + 2 (L i  − 1) + (L o  − 1) (4L i  − 2)

((L i  ≤ D op )|(L o  ≤ D ip )) & (L i  ≥ 4)

Pruned decomposed transform

OPERtot of (8) using Table 2

(L i  > D op ) & (L o  > D ip )

The selection of proper permutation/allocation structure directly relates to the considered above problem of selection of an optimal commutation-pruning implementation structure casted and treated as a combinational hypotheses testing task. All feasible hypotheses \( {\left\{{\mathrm{H}}_h\right\}}_{h=1}^{12} \) relate to formal implementation structures specified in Table 1. Now, we are ready to find the best permutation/allocation structure in the sense of the imposed quality measure (in our case in the sense of the lowest possible number of required arithmetical operations).

3.1 Analysis of the hypotheses

Let us analyze, first, the pruned decomposed transform and deduce whether the direct or recursive method would be preferable. The total number of arithmetic operations (OPERtot) required by the DFTDIF−DIT−Pr and the DFTDIT−DIF−Pr depends on the number of operations needed to be performed to implement the input stage (OPERinput), the output stage (OPERoutput), and the intermediate stage (D ip D op OPERDFTP), correspondingly. Thus, one could express OPERtot of both pruned decomposed transforms as
$$ \mathrm{O}\mathrm{P}\mathrm{E}{\mathrm{R}}_{\mathrm{tot}}=\mathrm{O}\mathrm{P}\mathrm{E}{\mathrm{R}}_{\mathrm{input}}+{D}_{ip}{D}_{op}\mathrm{O}\mathrm{P}\mathrm{E}{\mathrm{R}}_{\mathrm{DFTP}}+\mathrm{O}\mathrm{P}\mathrm{E}{\mathrm{R}}_{\mathrm{output}}. $$
(8)

According to (8), OPERtot depends on L i , L o , N, D ip , D op , and the algorithm employed to implement the D ip D op DFT P blocks (OPERDFTP).

At the input and output stages, there are multiplications by one, so those multiplications are avoided at all in our approach. Also, the multiplications by one at the input stage are also avoided depending on whether DFTDIF−DIT−Pr or DFTDIT−DIF−Pr was executed in the particular employed pruned decomposed transform modality.

If the DFTDIF−DIT−Pr modality is employed (see the general diagram in Fig. 1), then:
  • At the input stage, the multiplications by one are excluded when n 1 = n 2 = 0 and k 1 = 0.

  • Furthermore, the multiplications by one at the output stage are also avoided when n 1 = 0 or k 2 = 0.

Therefore, the DFTDIF−DIT−Pr modality always requires fewer complex multiplications to compute the output stage than the DFTDIT-DIF-Pr modality (this is reported in Tables 2 and 3).

On the other hand, if the DFTDIT-DIF-Pr modality is used (see Fig. 2), then:
  • At the input stage, the multiplications by one are excluded when n 2 = 0 or k 1 = 0.

  • Also, at the output stage, the multiplications by one are avoided when k 1 = k 2 = 0 or n 1 = 0.

Therefore, the DFTDIT-DIF-Pr modality always requires fewer complex multiplications at the input stage than the DFTDIF−DIT−Pr modality (as it is corroborated in the analysis reported in Tables 2 and 3).

The output stage of both pruned decomposed transform modalities can be computed by the direct addition of complex multiplications or a kind of recursive algorithm as those proposed in [23] (referred to as the 2BF filtering method), which reduces the number of required multiplications by about half. The number of arithmetic multiplications required by the output stage of the DFTDIF−DIT−Pr algorithm is equal to 4 (L o  − D ip ) (D op  − 1) when (L o  > D ip ) and (D op  < 4). Next, the number of arithmetic multiplications is equal to (L o  − D ip ) (2D op  + 2) when (L o  > D ip ) and (D op  ≥ 4). Thus, the 2BF filtering algorithm can be effectively used to compute the output stage.

On the other hand, the number of arithmetic multiplications required to compute the output stage of the DFTDIT-DIF-Pr algorithm is equal to 4(L o  − 1) (D op  − 1) when (D op  < 4); and the number of arithmetic multiplications is equal to (L o  − 1) (2D op  + 2) when (D op  ≥ 4). Thus, the 2BF filtering algorithm can also be effectively employed to compute the output stage.

In [23], it was proven that the 2BF filtering method is more efficient than the direct addition of complex multiplications when the number of input elements is larger than 4 (when the number of input elements is equal to 4, both methods manifest the same operational complexity performances). The output stages of both pruned decomposed transforms have the same structures, so same sort of commutations is required to efficiently compute the output stage of the DFTDIT−DIF−Pr. The expressions for OPERinput and OPERouput for the DFTDIF−DIT−Pr and the DFTDIT−DIF−Pr are listed in Table 2, where it is implicitly assumed that each complex multiplication requires six arithmetic operations (four real multiplications and two real additions), and each complex addition requires two arithmetic operations (two real additions).

The performances of the pruned decomposed transforms depend on the decomposition factors, D ip and D op . A simple analysis can be carried out to deduce which decomposition factors are preferable to be used. Our unified commutation-pruning method performs the decomposition of the DFT N into three stages of smaller dimension DFTs and pruning part of those inputs that are equal to zero and/or part of those outputs that are not needed to compute the final Fourier coefficients.

Thus, the decomposed transform algorithm always selects a pair (D ip , D op ) for which the largest DFTs could be successfully pruned, or equivalently, a pair (D ip , D op ) for which the intermediate stage results in the smallest dimension DFTs.

The DFTs of the intermediate stage have a size of N/D ip D op  ≡ P, so D ip and D op should be chosen as large as possible. Furthermore, the values for the decomposition factors should satisfy the bound N/D ip  ≥ L i (where, N/D ip must be close to but higher than L i ) and N/D op  ≈ L o , as it was considered in the derivation of (5). Hence, the pair of decomposition factors (D ip , D op ) closest to (N/L i , N/L o ) that satisfy D ip  ≤ N/L i are used by the decomposed transform algorithm, according to the proximity evaluated by its Euclidean distance.

Let us now consider the cases when the number of input elements (L i ) or the number of the required Fourier coefficients (L o ) is too small. In these cases, for the both modalities, the general diagrams presented in Figs. 2 and 1 clarify the following features of the DFTDIT−DIF−Pr and the DFTDIF−DIT−Pr algorithms, respectively.
  • If L i  ≤ D op , at most one input of each DFT P (i.e., the first one) in the intermediate stage would be applied; therefore, their P outputs would be replicas of that single input.

  • For L o  ≤ D ip , only the first output of each DFT P (this corresponds to a simple addition of the input elements) is required to compute the final Fourier coefficients.

Thus, inefficient implementations of the DFT P s yield the inequality-type constraints Li ≤ D op or L o  ≤ D ip . In these cases, our method commutes to efficiently perform the direct computation of the DFT N or an efficient recursive alternative (via performing the 2BF filtering technique).

Sorensen et al., in [23], proposed a method to compute a subset of the output components of their proposed specific DFT decomposition; this algorithm was referred to as a 2BF filtering method. The 2BF filtering method [23] was derived as a modification of the previously addressed Goertzel algorithm [25]. The 2BF filtering method takes advantages of the periodicity and the shifted cyclic convolution shape between the input sequence and the \( {W}_N^{nk}={e}^{-j\left(2\pi /N\right)kn} \) factor.

The transfer function H(z) of a system that performs the 2BF filtering method is given by the equation
$$ H(z)=\frac{z^{-1}\left(1-{z}^{-1}{W}_N^{-k}\right)}{1-2 \cos \left(\frac{2\pi k}{N}\right){z}^{-1}+{z}^{-2}} $$
(9)
The corresponding algorithmic diagram of the second-order 2BF method is presented in Fig. 4. Thus, (9) is the mathematical definition of H3.
Fig. 4

Algorithmic diagram of the second order 2BF filtering method

The poles of the system transfer function (the roots of the polynomial in the denominator of H(z)) have to be evaluated L times (n = 0, 1, 2,…, L − 1), while the zeros of the system transfer function (the roots of the numerator of H(z)) only once. Here, L represents the number of consecutive non-zero input elements of the 2BF filter; i.e., in the opposite case, it represents the number of consecutive non-zero output components of the employed pruned decomposed transform modality (DFTDIF−DIT−Pr or DFTDIT−DIF−Pr).

The computation of each pole of (9) requires two arithmetic multiplications (two real multiplications) and two arithmetic additions (two real additions). Furthermore, the computation of the zeros of (9) requires four arithmetic multiplications and four arithmetic additions only.

The Q 1 node in Fig. 4 is initialized with f(L − 1); therefore, the computation starts from n = L − 2. When n = 0, the complex addition of the input is only required; then, the zero is computed after such a delay. Such computational organization saves two arithmetic multiplications and six arithmetic additions for finding of each required output component.

The 2BF filtering method employed to compute the output components required by the pruned decomposed transform performed by the DFTCOMM−DIF−DIT−Pr or the DFTCOMM−DIT−DIF−Pr algorithm can be featured as the following multistage procedure:
  • The structure of the DFTDIF−DIT−Pr contains D ip sets of D op DFT P s from which the final outputs are computed (see the general diagram in Fig. 1).

  • The DFTCOMM−DIF−DIT−Pr algorithm employs the 2BF filtering method to implement the output stage of DFTDIF−DIT−Pr with L = L o , if ( (L i  > D op ) & (L o  > D ip ) ) & ( (L o  > D ip )&(D op  ≥ 4) ) (as featured in Tables 2 and 3). Here, the required arithmetic operations are specified as follows: the number of arithmetic multiplications are equal to NumArithMult 2BF  = (L o  − D ip )(2D op  + 2) and the number of arithmetic additions are equal to NumArithAdd 2BF  = 2 D ip (D op  − 1) + (L o  − D ip )(4D op  − 2).

  • Furthermore, the DFTCOMM−DIF−DIT−Pr algorithm employs the 2BF filtering method exclusively with L = L i , if ( (L i  ≤ D op ) | (L o  ≤ D ip ) ) & (L i  ≥ 4) (as featured in Table 3) to compute the required Fourier coefficients. Here, the required arithmetic operations are specified as follows: NumArithMult 2BF  = (L o  − 1)(2L i  + 2) and NumArithAdd 2BF  = 2(L i  − 1) + (L o  − 1)(4L i  − 2).

In contrast, the DFTCOMM-DIT-DIF-Pr algorithm differs from the abovementioned in the following features:
  • The structure of the DFTDIT-DIF-Pr contains D op sets of D ip DFT P s from which the final outputs are computed (as featured in Fig. 2).

  • The DFTCOMM-DIT-DIF-Pr algorithm employs the 2BF filtering method to implement the output stage of DFTDIT-DIF-Pr with L = L o , if ( (L i  > D op ) & (L o  > D ip ) ) & (D op  ≥ 4), (as featured in Tables 2 and 3). Here, the required arithmetic operations are specified as follows: NumArithMult 2BF  = (L o  − 1)(2D op  + 2), and NumArithAdd 2BF  = 2 (D op  − 1) + (L o  − 1)(4D op  − 2).

  • On the other hand, the DFTCOMM-DIT-DIF-Pr algorithm employs the 2BF filtering method exclusively with L = L i , if ( (L i  ≤ D op ) | (L o  ≤ D ip ) ) & (L i  ≥ 4) (as reported in Table 3) to compute the required Fourier coefficients. Here, the required arithmetic operations are specified as follows: NumArithMult 2BF  = (L o  − 1)(2L i  + 2) and NumArithAdd 2BF  = 2(L i  − 1) + (L o  − 1)(4L i  − 2).

The computation of each input and/or output element in both cases detailed above is executed according to the diagram presented in Fig. 4. In closing, we note that the pseudo-code presented in the Appendix (see Fig. 9) contains all scripts needed to compute each Fourier coefficient employing the 2BF filtering method.

Note once again that the 2BF filtering method has to be employed if L i is larger or equal to 4, in which case, it manifests a higher efficiency than the direct method for computing the DFT N in (1). The total number of arithmetic operations required by our proposed method is reported in Table 3.

3.2 Selection of the permutation/allocation structure

In Fig. 5, the total number of required arithmetic operations to compute the DFTDIF−DIT−Pr from [24] (H9), DFTCOMM−DIF−DIT−Pr (H11), and DFTCOMM−DIT−DIF−Pr (H12) modalities are plotted for different values of L i and L o for the test examples with N = 8192 and N = 6561 (It is assumed that the DFT P s are implemented employing the split-radix algorithm from [26] for N = 8192 and employing the radix-3 algorithm from [27] for N = 6561.) All the competing alternatives corresponding to three feasible arrangements (H9, H11, and H12) in the considered permutation/allocation structure are featured in Fig. 5. The DFTDIF−DIT−Pr or the DFTDIT−DIF−Pr could be used to implement the pruned decomposed transform in the DFTCOMM−DIF−DIT−Pr and DFTCOMM−DIT−DIF−Pr techniques. Here, the D ip and D op values are the pair specified by the rough selection method (the proximity evaluated by its Euclidean distance is referred as roughDP) and those obtained by an exhaustive search method (the total numbers of operations required to implement the DFTDIF−DIT−Pr and the DFTDIT−DIF−Pr were evaluated for each possible pair of (D ip , D op ), and, then, the pair (D ip , D op ) with the best performance metric is selected; this selection method is referred as exhDP). Fig. 5a–f demonstrate that two commutation-pruning techniques (related to hypotheses H11 and H12) require the same or smaller number of arithmetic operations than that specified by hypothesis H9. Next, it is necessary to make a choice between H11 and H12.
Fig. 5

Number of arithmetic operations required to compute the DFTDIF-DIT_Pr, the DFTCOMM−DIF−DIT−Pr, and the DFTCOMM−DIT−DIF−Pr: a for a constant value of L i and different tested values of L o  = {1, 2,…, N} when N = 8192; b for a constant value of L o and different tested values of L i  = {1, 2,…, N} when N = 6561; c for a constant value of L o and different tested values of L i  = {1, 2,…, N} when N = 8192; d for a constant value of L i and different tested values of L o  = {1, 2,…, N} when N = 6561; e zoom of Fig. 5a; and f zoom of Fig. 5c

Graphs in Fig. 5 indicate that the number of operations required to perform our commutation-pruning technique (DFTCOMM−DIF−DIT−Pr and DFTCOMM−DIT−DIF−Pr) with the selected decomposition factors using the roughDP method are equal to or slighty greater than those, in which the decomposition factors are specified employing exhDP. The differences correspond to the regions where the commutation conditions prescribe performing the pruned decomposed transform instead of the 2BF filtering method.

The DFTDIT−DIF−Pr modality requires the same or a smaller number of arithmetic operations than the competing DFTDIF−DIT−Pr for all the cases where the pruned decomposed transform is performed (as it follows from the data reported in Fig. 5). Since the same decomposition factors (D ip , D op ) are used in both pruned decomposed transforms, it is sufficient to compare the number of required operations by their input and output stages (OPERinput + OPERoutput) reported in Table 2 to distinguish which one is the most efficient. The comparison for the cases L i  ≤ D op and L o  ≤ D ip is not needed since in such scenarios, a direct or recursive method is employed instead of a pruned decomposed transform. For scenarios with D op  < 4, both pruned decomposed transforms require the same number of arithmetic operations for their execution. Otherwise, for D op  ≥ 4, the execution of DFTDIF−DIT−Pr requires 2D ip D op  − 8D ip  − 2D op  + 8 more arithmetic operations than DFTDIT−DIF−Pr demonstrating that the latter manifests always the same or a better performance. Thus, from the combinational permutation analysis, it follows that it is always desirable to perform the DFTDIT−DIF−Pr when a pruned decomposed transform would be required. In the following section, an efficient implementation of that proposed unified commutation-pruning technique is detailed considering that the pruned decomposed transform is implemented using the DFTDIT−DIF−Pr. In summary, we now resume that the performed combinational hypothesis testing-based optimal selection of the preferable computational structure of the decomposed DFTs made the decision in favor of hypothesis H12; this yields the proposed DFTCOMM−DIT−DIF−Pr method (referred further on for simplicity as DFTCOMM) with the highest possible computational efficiency. Being the optimal decision of the performed “brute force search” based testing of all feasible hypotheses, this method is guaranteed to be globally optimal one and thus is strongly recommended for performing the required commuting between three techniques to implement the overall composite DFT in the following arrangement mode: the direct method, the recursive method, and the pruned decomposed transform implemented via DFTDIT−DIF−Pr.

4 Comparison with other competing algorithms

A variety of competing methods for pruning the DFTs in arbitrary (non-sparse) computational scenarios have been addressed in the literature (see [1, 3, 1319, 23, 24]). In [24], the FFTDIF−DIT−TD modality (that we here refer to as DFTDIF−DIT−Pr) was proposed as an alternative technique for pruning the input and/or the output of DFTs. That method [24] was compared with other pruning techniques reported in the literature until 2009. Comparisons of the methods proposed by Bouguezel et al. [15], Fan et al. [16], Sreenivas et al. [17], Roche [18], and the DFTDIF−DIT−Pr reported in [24] demonstrated that the DFTDIF−DIT−Pr modality requires fewer arithmetic operations than those of [1517], while attaining the operational performances similar to that of [18]. Additionally, in Section 3, it was corroborated that our proposed DFTCOMM technique requires equal or less arithmetic operations than [24]. Here beneath, we compare our approach with the recently reported most prominent competing pruning methods.

4.1 Comparisons with pruning-based algorithms

The first competing algorithm for pruning the output of a SRFFT was reported in [3]. That so-called SRFFTpruning algorithm was developed for an implicit restriction that only a few consecutive output components (a number L equal to a power of two) are required. Fig. 6 reports the number of arithmetic operations required to perform SRFFTpruning in comparison with our unified DFTCOMM method for multiple output pruning examples using the decomposition factors (D ip , D op ) evaluated via the roughDP method and those specified by the exhDP method, respectively.
Fig. 6

Number of arithmetic operations required to perform the DFTCOMM, SRFFT(noprun), and the SRFFTpruning algorithms; parameters D ip and D op are selected using roughDP and exhDP methods for a constant value of L i , and different tested values of L o  = {21, 22,…, N}: a for N = 262,144 and L i  = {33, 1027, 262,144}; b for N = 1024 and L i  = {13, 90, 1024}

In both cases, it is considered that the DFTs of length P required by the intermediate stage of the pruned decomposed transform have been implemented by applying the split-radix FFT, e.g., [26]. Therefore, the total number of arithmetic operations required by our proposed DFTCOMM method in comparison with the competing pruning-based algorithms can be found in Table 4. The savings in the number of arithmetic operations attained with the new developed DFTCOMM technique are reported in Tables 5 and 6.
Table 4

Total number of arithmetic operations required to compute the SRFFTpruning, SRFFTpruning-time-shift, and DFTCOMM algorithms

Algorithm

Number of required real arithmetic operations

Conditions

SRFFTpruning

\( 6\left\{{\displaystyle \sum_{k=1}^d\left[{N_B}_{(k)}{N_W}_{(k)}\right]}\right.+\left.\frac{1}{2}\left({N_B}_{\left(d+1\right)}{N_W}_{\left(d+1\right)}\right)\right\}+2\left\{N\cdot d+{\displaystyle \sum_{k=0}^{r-\left(d+1\right)}\left[L{2}^k\right]}\right\} \)

SRFFTpruning-time-shift

\( 6\left\{{\displaystyle \sum_{k=1}^{r-d}\left[{N_B}_{(k)}2L\right]+}\right.\left.{\displaystyle \sum_{k=r-d+1}^{r-1}\left\{{N_B}_{(k)}\left[{N_W}_{(k)}+2\right]\right\}}\right\}+2\left\{N\left(d-1\right)+N\right\} \)

DFTCOMM

6(L o  − 1)(L i  − 1) + 2L o (L i  − 1)

{(L i  ≤ D op )|(L o  ≤ D ip )} & (L i  < 4)

 

(L o  − 1)(2L i  + 2) + 2(L i  − 1) + (L o  − 1)(4L i  − 2)

{(L i  ≤ D op )|(L o  ≤ D ip )} & (L i  ≥ 4)

 

6(L i  − D op )(D ip  − 1) + D ip D op (4P log2(P) − 6P + 8) + 6(L o  − 1)(D op  − 1) + 2L o (D op  − 1)

{(L i  > D op ) & (L o  > D ip )} & (D op  < 4)

 

6(L i  − D op )(D ip  − 1) + D ip D op (4P log2(P) − 6P + 8) + (L o  − 1)(2D op  + 2) + 2(D op  − 1) +

{(L i  > D op ) & (L o  > D ip )} & (D op  ≥ 4)

Where:

• N → Length of the full transform

• r = log2(N) → Number of stages

• L = 2 d  → Number of consecutive outputs that should be computed for SRFFTpruning algorithm or number of consecutive non-zero inputs for SRFFTpruning-time-shift algorithm

• k → Stage k

\( {N}_{B(k)}=\frac{2^{k-1}+{\left(-1\right)}^k}{3}\to k=1,2,\dots, r\to \) Number of twiddle factor blocks

• N W(k) = 2(2 r − k  − 1) → k = 1, 2, …, r → Number of twiddle factors for each block

• L i  → Number of consecutive input elements different from zero

• L o  → Number of consecutive output that should be computed

• D ip and D op  → Integer decomposition factors

• N/D ip D op  ≡ P

Table 5

Savings in the number of arithmetic operations attained with the DFTCOMM algorithm in comparison with the competing SRFFT (noprun) and SRFFTpruning methods for N = 262,144

DFTCOMM in comparison with:

L o

L i

Savings

SRFFT(noprun)

{21,22,…, N}

N = 218

DFTCOMM, 42.76 % with output pruning

SRFFTpruning

  

DFTCOMM, 2.96 % with output pruning

SRFFT(noprun)

 

1027

DFTCOMM, 75.02 % with input-output pruning at the same time

SRFFTpruning

  

SRFFTpruning fails to deliver a result

SRFFT(noprun)

 

33

DFTCOMM, 91.35 % with input-output pruning at the same time

SRFFTpruning

  

SRFFTpruning fails to deliver a result

Table 6

Savings in the number of arithmetic operations attained with the DFTCOMM algorithm in comparison with the competing SRFFT (noprun) and SRFFTpruning methods for N = 1024

DFTCOMM in comparison with:

L o

L i

Savings

SRFFT(noprun)

{21,22,…, N}

N = 210

DFTCOMM, 36.48 % with output pruning

SRFFTpruning

  

DFTCOMM, 2.73 % with output pruning

SRFFT(noprun)

 

90

DFTCOMM, 59.30 % with input-output pruning at the same time

SRFFTpruning

  

SRFFTpruning fails to deliver a result

SRFFT(noprun)

 

13

DFTCOMM, 81.65 % with input-output pruning at the same time

SRFFTpruning

  

SRFFTpruning fails to deliver a result

From Fig. 6, one can deduce that our proposed DFTCOMM method requires fewer arithmetic operations than the competing SRFFTpruning method in almost all the test cases (with the only one exception for the case L o  = N/2 and L o  = N/4). Next, Tables 5 and 6 report the savings in the number of arithmetic operations attained with our DFTCOMM in comparison with the competing SRFFT and the SRFFTpruning techniques. In the scenarios with L o  = N and L i  = {21, 22,…, N}, the DFTCOMM algorithm manifests 2.96 and 2.73 % savings in the number of arithmetic operations in comparison with the SRFFTpruning for N = {262,144, 1024}, respectively.

In other cases, from Table 5, it follows that in the scenarios with L i  = 1027, L i  = 33, and L o  = {21, 22,…, N}, the SRFFTpruning method fails to deliver a result at all. Thus, from Table 5, it follows that in the cases when L i  = N = 262,144, L i  = 1027, L i  = 33, and L o  = {21, 22,…, N}, the DFTCOMM algorithm produces savings of 42.76, 75.02, and 91.35 %, respectively, in the number of arithmetic operations required to compute the composite length DFT in comparison with the competing SRFFT algorithm. Furthermore, from Table 6, it follows that in the scenarios with L i  = 90, L i  = 13, and L o  = {21, 22,…, N}, the SRFFTpruning method fails to deliver a result at all. Thus, from Table 6, it follows that in the cases when L i  = N = 1024, L i  = 90, L i  = 13, and L o  = {21, 22,…, N}, the DFTCOMM algorithm produces savings of 36.48, 59.30, and 81.65 %, respectively, in the number of arithmetic operations required to compute the composite length DFT in comparison with the competing SRFFT algorithm.

Yuan et al., in [14], proposed another competing, the so-called SRFFTpruning−time−shift method via modifying the SRFFTpruning employing a time shifting approach that yields the input pruning algorithm based on the SRFFT methodology for L consecutive non-zero input elements. It is noteworthy to stress that the SRFFTpruning−time−shift approach implicitly assumes that lengths L and N may take values equal to the power of two only.

Figure 7 reports the number of required arithmetic operations to execute our proposed unified DFTCOMM method and those required by the competing pruned DFTs of [14]. These results verify that our approach requires fewer arithmetic operations than those required to perform the SRFFTpruning−time−shift algorithm in all the reported tests. Again, it is implicitly assumed that the DFTs of length P involved in the DFTDIT−DIF−Pr used by our DFTCOMM have been computed using the split-radix FFT [26], as reported in Table 4.
Fig. 7

Number of arithmetic operations required to perform the DFTCOMM, SRFFT(noprun), and the SRFFTpruning algorithms; parameters D ip and D op are selected using roughDP and exhDP methods for a constant value of L o , and different tested values of L i  = {21, 22, …, N}: a for N = 262,144 and L o  = {33, 1027, 262,144}; b for N = 1024 and L o  = {13, 90, 1024}

Next, Tables 7 and 8 report the savings in the number of arithmetic operations attained with our DFTCOMM in comparison with the competing SRFFT and the SRFFTpruning−time−shift techniques. In the scenarios with L o  = N and L i  = {21, 22,…, N}, the DFTCOMM algorithm manifests 5.11 and 8.71 % savings in the number of arithmetic operations in comparison with the SRFFTpruning−time−shift for N = {262,144, 1024}, respectively.
Table 7

Savings in the number of arithmetic operations attained with the DFTCOMM algorithm in comparison with the competing SRFFT (noprun) and SRFFTpruning-time-shift methods for N = 262,144

DFTCOMM in comparison with:

L i

L o

Saving

SRFFT(noprun)

{21,22,…, N}

N = 218

DFTCOMM, 43.26 % with input pruning

SRFFTpruning-time-shift

  

DFTCOMM, 5.11 % with input pruning

SRFFT(noprun)

 

1027

DFTCOMM, 76.24 % with input-output pruning at the same time

SRFFTpruning-time-shift

  

SRFFTpruning-time-shift fails to deliver a result

SRFFT(noprun)

 

33

DFTCOMM, 92.11 % with input-output pruning at the same time

SRFFTpruning-time-shift

  

SRFFTpruning-time-shift fails to deliver a result

Table 8

Savings in the number of arithmetic operations attained with the DFTCOMM algorithm in comparison with the competing SRFFT (noprun) and SRFFTpruning-time-shift methods for N = 1024

DFTCOMM in comparison with:

L i

L o

Saving

SRFFT(noprun)

{21,22,…, N}

N = 210

DFTCOMM, 38.22 % with input pruning

SRFFTpruning-time-shift

  

DFTCOMM, 8.71 % with input pruning

SRFFT(noprun)

 

90

DFTCOMM, 59.22 % with input-output pruning at the same time

SRFFTpruning-time-shift

  

SRFFTpruning-time-shift fails to deliver a result

SRFFT(noprun)

 

13

DFTCOMM, 82.45 % with input-output pruning at the same time

SRFFTpruning-time-shift

  

SRFFTpruning-time-shift fails to deliver a result

In other test cases, from Tables 7 and 8, it follows that for L o  = {1027, 90}, L o  = {33, 13}, and L i  = {21, 22,…, N}, the SRFFTpruning−time−shift algorithm fails to deliver a result at all. Furthermore, from Table 7, it follows that in the scenarios with L o  = {N, 1027, 33} and L i  = {21, 22,…, N}, our DFTCOMM attains 43.26, 76.24, and 92.11 % savings for N = 262,144, respectively, in the number of arithmetic operations required to compute the composite length DFT. In addition, from Table 8, it follows that in the scenarios with L o  = {N, 90, 13} and L i  = {21, 22,…, N}, our DFTCOMM attains 38.22, 59.22, and 82.45 % savings for N = 1024, respectively, in the number of arithmetic operations required to compute the composite length DFT.

Note that our DFTCOMM always requires fewer arithmetic operations than the competing SRFFTpruning and SRFFTpruning−time−shift algorithms due to the different butterfly schemes employed to implement the split-radix FFT algorithms [26] and the unified commutation-pruning technique employed (see Section 3). The SRFFTpruning and SRFFTpruning−time−shift algorithms perform the two-butterfly scheme [26], while our DFTDIT−DIF−Pr algorithm employs the three-butterfly scheme to achieve a reduction in the number of arithmetic operations required to implement the DFT P blocks. Furthermore, graphs of Fig. 6 report that the SRFFTpruning algorithm fail to deliver a result at all in the scenarios with L equal to N due to their algorithmic construction as reported by the authors of [3]. For this reason, this algorithm cannot present a valid value for the last test of L o (it is simply unable to stop to prune at all). In addition, Fig. 6 reports minimal differences between the numbers of arithmetic operations attained by the DFTCOMM evaluated using the roughDP- or exhDP-based selection for specifying D ip and D op . In summary, the number of arithmetic operations required to compute the SRFFTpruning, SRFFTpruning-time-shift, and DFTCOMM algorithms can be found in Table 4.

4.2 Comparison with the SFFT-related algorithms

In a context of pruned DFTs, real-world sensing scenarios are characterized by the uncertainties attributed to zero-padded input data acquisition modes with variable composite length windowing of the input and/or output Fourier transform sequences, in general cases, with non-sparse Fourier spectrum [1012]. In contrast, the celebrated SFFT method developed and featured in [20] presumes “sparsity” of the Fourier spectrum that requires that majority of the Fourier coefficients are zeros or negligible; e.g., the authors of [20] exemplified such sparsity level at approximately 89 %, i.e., up to 89 % of the Fourier transform coefficients are to be zeroes or negligible for operability of their SFFT. Otherwise, the DFT should be specified and treated as a non-sparse transform.

Currently, a family of novel efficient algorithms for computing the FFTs applicable for sparse sensing scenarios when only a few Fourier transform coefficients (k s largest coefficients of the N-length Fourier transform) of the input signal x are different from zero have been developed [20, 21], which compose a family of the so-called SFFT methods. To compute a reliable SFFT for typical high N > 210, the sparsity level constraint requires that majority of the Fourier coefficients are zeros [20] (or negligible to be discarded). Such model assumptions are valid, for example, in video compressing applications [20]. Therefore, if majority of the Fourier transform coefficients are supposed to be zeros or can be discarded, then efficient computing techniques from the SFFT family can be employed. The celebrated algorithms from such a family are the SFFTv1 and the SFFTv2 developed and featured in [20] where the sparsity level was exemplified at 89 % of zero (negligible) Fourier coefficients. In [21], the SFFTv3 and SFFTv4 algorithms were proposed, where some computational improvements were introduced. SFFTv3 was implemented in [28] while the program code for implementation of the SFFTv4 algorithm is not available at this time. Another competing technique for computing of the FFT of sparse (in the frequency domain) signals was addressed in [22] as the so-called FADFT-2 algorithm from the AAFFT library [22]. However, in [20, 21], it was corroborated that the SFFT-related algorithms manifest better operational performances than FADFT-2 of [22].

To perform valid test comparisons between the SFFTv1, SFFTv2, SFFTv3, and the DFTCOMM algorithms, those should be tested under the same conditions and constrains. Here, we use the following feasible constraints: the values of N vary as follows: N = {26, 27,…, 220} and k s = L o , where L o represents the number of consecutive output coefficients to be calculated. In different test scenarios, the SFFTv1, SFFTv2, and SFFTv3 algorithms deliver successful results: the first of them for N = {213, 214,…, 220} and k s = L o  = 50, the second of them for N = {213, 214,…, 220} and k s = L o  = 50, and finally, the third of them for N = {210, 211,…, 220} and k s = L o  = 50, respectively. Furthermore, it was experimentally corroborated that the DFTCOMM algorithm was able to deliver efficient results in all such tested sparse scenarios, as reported in Table 9.
Table 9

Comparisons of the SFFTv1, SFFTv2, SFFTv3, and DFTCOMM algorithms for different sizes (N) of the signal x, with N = L i  = {26, 27,…, 220} and k s = L o  = 50

L i  = N

k s = L o

SFFTv1

SFFTv2

SFFTv3

DFTCOMM

26 = 64

50

*

*

*

27 = 128

50

*

*

*

28 = 256

50

*

*

*

29 = 512

50

*

*

*

210 = 1024

50

*

*

211 = 2048

50

*

*

212 = 4096

50

*

*

213 = 8192

50

214 = 16,384

50

215 = 32,768

50

216 = 65,536

50

217 = 131,072

50

218 = 262,144

50

219 = 524,288

50

220 = 1,048,576

50

(*) The program execution is aborted

() The program execution is successful

In addition, DFT computations for other sparse test scenarios with different values of N and k s were run, in particular, for N = L i  = {213, 214,…, 217} and k s = L o  = {1, 2, …, k smax} with k smax = 11 % of N. The test scenarios for the SFFT algorithms delivered successful results only for a few tested values of k s. For example, the SFFTv1 algorithm is executed successfully for N = {213, 215} and k s = {1, 2,…, 50}, for N = 214 and k s = {1, 2, …, 50}  {56, 57, …, 63}, for N = 216 and k s = {1, 2, …, 50}  {64, 65, …, 97}, and for N = 217 and k s = {1, 2,…, 74}.

The SFFTv2 algorithm is executed successfully for N = {213, 214,…, 217} and k s = {1, 2,…, 50}, while, the SFFTv3 algorithm performed successfully for N = 213 and k s = {4, 5,…, 673}, for N = 214 and k s = {4, 5,…, 1346}, for N = 215 and k s = {4, 5,…, 2692}, for N = 216 and k s = {4, 5,…, 5385}, and for N = 217 and k s = {4, 5,…, 10,771}. Furthermore, the DFTCOMM algorithm is executed successfully for all test cases (for N = {213, 214,…, 217} in combination with all k s = {1, 5,…, k smax}, as follows from the data reported in Table 10.
Table 10

Comparisons of the SFFTv1, SFFTv2, SFFTv3, and DFTCOMM algorithms for different sizes (N) of the signal x, with N = L i  = {213, 214,…, 217} and k s = L o  = {1, 2,…, k smax} in the tested sparse scenarios with k smax ~ 11 % of N

L i  = N

1 ≤ k s ≤ k smax

SFFTv1

213

1 ≤ k s ≤ 901

k s = {1, 2,…, 50}

214

1 ≤ k s ≤ 1802

k s  = {1, 2, …, 50}  {56, 57, …, 63}

215

1 ≤ k s ≤ 3604

k s = {1, 2,…, 50}

216

1 ≤ k s ≤ 7208

k s = {1, 2, …, 50}  {64, 65, …, 97}

217

1 ≤ k s ≤ 14,417

k s = {1, 2,…, 74}

L i =  N

1  k s k smax

SFFTv2

213

1 ≤ k s ≤ 901

k s = {1, 2,…, 50}

214

1 ≤ k s ≤ 1802

k s = {1, 2,…, 50}

215

1 ≤ k s ≤ 3604

k s = {1, 2,…, 50}

216

1 ≤ k s ≤ 7208

k s = {1, 2,…, 50}

217

1 ≤ k s ≤ 14,417

k s = {1, 2,…, 50}

L i =N

1≤  k s k smax

SFFTv3

213

1 ≤ k s ≤ 901

k s = {4, 5,…, 673}

214

1 ≤ k s ≤ 1802

k s = {4, 5,…, 1346}

215

1 ≤ k s ≤ 3604

k s = {4, 5,…, 2692}

216

1 ≤ k s ≤ 7208

k s = {4, 5,…, 5385}

217

1 ≤ k s ≤ 14,417

k s = {4, 5,…, 10,771}

L i =  N

1k s k smax

DFT COMM

213

1 ≤ k s ≤ 901

k s = {1, 2,…, k smax}

214

1 ≤ k s ≤ 1802

k s = {1, 2,…, k smax}

215

1 ≤ k s ≤ 3604

k s = {1, 2,…, k smax}

216

1 ≤ k s ≤ 7208

k s = {1, 2,…, k smax}

217

1 ≤ k s ≤ 14,417

k s = {1, 2,…, k smax}

Table 11 reports the absolute average errors attained with the SFFTv1, SFFTv2, SFFTv3, and DFTCOMM algorithms, for N = {213, 214,…, 218} and k s = L o  = 50. In all test cases, the FFTW algorithm from [29] was used as a reference for computing the absolute error measures.
Table 11

Average absolute errors attained in sparse scenarios with the SFFTv1, SFFTv2, SFFTv3, and DFTCOMM algorithms for N = {213, 214,…, 218} and k s = L o  = 50

L i  = N

k s = L o

SFFTv1

SFFTv2

SFFTv3

DFTCOMM

AbsError

AbsError

AbsError

AbsError

213

50

5.6162 × 10−5

5.0689 × 10−5

2.4973 × 10−5

2.7642 × 10−10

214

50

7.0000 × 10−4

6.3012 × 10−4

4.9943 × 10−5

1.8228 × 10−9

215

50

2.8526 × 10−4

2.5407 × 10−4

9.9883 × 10−5

1.6704 × 10−8

216

50

3.7305 × 10−4

3.6885 × 10−4

1.9976 × 10−4

1.5567 × 10−7

217

50

4.9801 × 10−7

4.8631 × 10−7

1.5437 × 10−7

2.6652 × 10−10

218

50

0.0023

0.0023

7.9905 × 10−4

8.2515 × 10−6

From the data reported in Table 11, it follows that for N = 8192 and k s = L o  = 50, the SFFTv1 and SFFTv2 algorithms manifest very close absolute error values; in particular, the attained average absolute error values were 5.6162 × 10−5 and 5.0689 × 10−5, respectively. However, the SFFTv3 attains a lower absolute average error values than other SFFT versions. It is noteworthy to mention that the lowest absolute average error was attained with the DFTCOMM algorithm at a value of 2.7642 × 10−10.

In addition, Fig. 8 reports the absolute values of errors of the compared tested SFFTv3 and the DFTCOMM algorithms for N = 8192 and k s = L o  = 50 under the same sparse computing scenarios.
Fig. 8

Measures of absolute error values attained with the SFFTv3 and DFTCOMM algorithms in sparse scenarios for N = 8192 and k s = L o  = 50: a SFFTv3 and DFTCOMM; b DFTCOMM

On the other hand, the SFFT-related algorithms demonstrate reliable operation for specific input parameter combinations, i.e., they are dependent on the combination of the dimension N of the input signal x, and the sparsity factor k s. In contrast, the DFTCOMM algorithm manifests the operational robustness in the sense that it does not subject to any of such dimensional limitation and demonstrated perfect operational performances in all tested harsh (non-sparse) computational scenarios. Furthermore, all SFFT-related algorithms are probabilistic-type techniques [20, 21], in which the desired k s largest coefficients of the Fourier spectrum of the input sequence are reconstructed (approximated) with a high probability (not mandatory with probability one). In contrast, the DFTCOMM algorithm is a deterministic technique, and it produces more reliable and accurate results than the family of the SFFT-related algorithms (as demonstrated in Fig. 8 and Tables 10 and 11).

It is also worthwhile to note that presently (in the sparsity-guaranteed computational scenarios only), the SFFT-related algorithms outperform the DFTCOMM in the computational speed due to their specially devised execution parallelism [20, 21, 28]. From the family of the SFFT-related algorithms, the SFFTv3 [28] manifests the most speed-up computational performances for any input sequence dimension N and any feasible value k s in the sparsity-guaranteed scenarios only; in particular, when approximately only 8.2 % (or lower number) of the Fourier coefficients of the input signal are significant, thus not discarded (as shown in Table 10). In contrast, in all comparable (sparse or non-sparse) computational scenarios, the DFTCOMM algorithm manifested superior accuracy performances (lower absolute error values) than those attained with the SFFT-related algorithms.

In closing, it is noteworthy to mention that in a majority of practical computational scenarios, the savings in the number of arithmetic operations achievable with the optimized unified DFTCOMM technique are significant. As a concluding example, refer to the test scenario with N = 8192 and L i  = L o =307 in which case the savings in the total number of required arithmetic operations attainable with the DFTCOMM algorithm in comparison with the most prominent competing split-radix FFT algorithm [3, 14, 23, 24] constitute 45 %.

5 Conclusions

We have developed a new technique that carries out an efficient computation of the DFTs of composite lengths of the input and/or output data sequences smaller than the dimension N of the full DFT/FFT. The addressed methodology unifies the commuting, filtering, and pruning paradigms yielding the new DFTCOMM method that outperforms the existing competing pruning-decomposition-based techniques in the sense of attainable savings in the number of required arithmetic operations.

Furthermore, our DFTCOMM method admits computing the DFT P blocks at the intermediate stage of the pruned decomposed transform using any existing FFT algorithm. Based on the performed treatment of the combinational hypotheses testing-type problem regarding all feasible allocation-pruning modalities, the decision in favor of the preferable hypothesis was made that yields the proposed DFTCOMM method. Being the globally optimal decision making result of testing the complete list of all feasible hypotheses, the DFTCOMM method guarantees to require a fewer or at most the same number of arithmetic operations for its execution than any other of the competing pruning-decomposition-based methods reported in the literature.

In addition, we have corroborated that, in the scenarios with non-guaranteed sparsity of the data Fourier spectra, the DFTCOMM method manifests better reliability and accuracy than the family of the celebrated competing SFFT-related algorithms; while in scenarios with severe Fourier spectrum non-sparsity (i.e., when the majority of the data Fourier spectrum coefficients take non-zero values, thus cannot be discarded), the DFTCOMM technique always outperforms the celebrated SFFT-related algorithms because all those simply fail to execute the program code in such uncertain computational scenarios.

Declarations

Acknowledgements

The authors would like to thank the anonymous reviewers for their constructive criticism and comments that helped to improve the presentation of the paper.

Open AccessThis article is distributed under the terms of the Creative Commons Attribution 4.0 International License (http://creativecommons.org/licenses/by/4.0/), which permits unrestricted use, distribution, and reproduction in any medium, provided you give appropriate credit to the original author(s) and the source, provide a link to the Creative Commons license, and indicate if changes were made.

Authors’ Affiliations

(1)
Department of Electrical-Electronic Engineering, Culiacan Technological Institute
(2)
Telecommunications Group, CINVESTAV-IPN Campus Guadalajara

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Copyright

© Castro-Palazuelos et al. 2015