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Joint optimization of MIMO radar waveform and biased estimator with prior information in the presence of clutter
EURASIP Journal on Advances in Signal Processing volume 2011, Article number: 15 (2011)
Abstract
In this article, we consider the problem of joint optimization of multiinput multioutput (MIMO) radar waveform and biased estimator with prior information on targets of interest in the presence of signaldependent noise. A novel constrained biased CramerRao bound (CRB) based method is proposed to optimize the waveform covariance matrix (WCM) and biased estimator such that the performance of parameter estimation can be improved. Under a simplifying assumption, the resultant nonlinear optimization problem is solved resorting to a convex relaxation that belongs to the semidefinite programming (SDP) class. An optimal solution of the initial problem is then constructed through a suitable approximation to an optimal solution of the relaxed one (in a least squares (LS) sense). Numerical results show that the performance of parameter estimation can be improved considerably by the proposed method compared to uncorrelated waveforms.
1 Introduction
Multiinput multioutput (MIMO) radar has attracted more and more attention recently [1–19]. Unlike the traditional phasedarray radar which can only transmit scaled versions of a single waveform, MIMO radar can use multiple transmitting elements to transmit arbitrary waveforms. Two categories of MIMO radar systems can be classified by the configuration of the transmitting and receiving antennas: (1) MIMO radar with widely separated antennas (see, e.g., [1, 2]), and (2) MIMO radar with colocated antennas (see, e.g., [3]). For MIMO radar with widely separated antennas, the transmitting and receiving elements are widely spaced such that each views a different aspect of the target. This type of MIMO radar can exploit the spatial diversity to overcome performance degradations caused by target scintillations [2]. In contrast, MIMO radar with colocated antennas, the elements of which in transmitting and receiving arrays are close enough such that the target radar cross sections (RCS) observed by MIMO radar are identical, can be used to increase the spatial resolution. Accordingly, it has several advantages over its phased array counterpart, including improved parameter identifiability [4, 5], and more flexibility for transmit beampattern design [6–19]. In this article, we focus on MIMO radar with colocated antennas.
One of the most interesting research topics on both types of MIMO radar is the waveform optimization, which has been studied in [6–19]. According to the target model used in the problem of waveform design, the current design methods can be divided into two categories: (1) point targetbased design [6–12], and (2) extended targetbased design [13–19]. In the case of point targets, the corresponding methods optimize the waveform covariance matrix (WCM) [6–8] or the radar ambiguity function [9–12]. The methods of optimizing the WCM only consider the spatial domain characteristics of the transmitted signals, while the one of optimizing the radar ambiguity function treat the spatial, range, and Doppler domain characteristics jointly. In the case of extended targets, some prior information on the target and noise are used to design the transmitted waveforms.
In [7], based on the CramerRao bound (CRB), the problem of MIMO radar waveform design for parameter estimation of point targets has been investigated under the assumption that the received signals do not include the clutter which depends on the transmitted waveforms. However, it is known that the received data is generally contaminated by the clutter in many applications (see, e.g., [13, 14]). It is noted that the CRB provides a lower bound on the variance when any unbiased estimator is used without employing any prior information. In fact, some prior information may be available in many array signal processing fields (see, e.g., [20–22]), which can be regarded as a constraint on the estimated parameter space. A variant of the CRB for this kind of the constrained estimation problem was developed in [20, 22], which is called the constrained CRB. Moreover, a biased estimator can lower the resulting variance obtained by any unbiased estimator generally [23–28]. The variant of the CRB for this case is named as the biased CRB. Furthermore, the variance produced by any unbiased estimator can be lowered obviously while both biased estimator and prior information are used. A variant of the CRB for this case was studied in [29], which can be referred to as the constrained biased CRB. Consequently, from the parameter estimation point of view, it is worth studying the waveform optimization problem in the presence of clutter by employing both the biased estimator and prior information.
In this article, we consider the problem of joint optimization of the WCM and biased estimator with prior information on targets of interest in the presence of clutter. Under the weighted or spectral norm constraint on the bias gradient matrix of the biased estimator, a novel constrained biased CRBbased method is proposed to optimize the WCM and biased estimator such that the performance of parameter estimation can be improved. The joint WCM and biased estimator design is formulated in terms of a rather complicated nonlinear optimization problem, which cannot be easily solved by convex optimization methods [30–32]. Under a simplifying assumption, this problem is solved resorting to a convex relaxation that belongs to the semidefinite programming (SDP) class [31]. An optimal solution of the initial joint optimization problem is then constructed through a suitable approximation to an optimal solution of the relaxed one (in a least squares (LS) sense).
The rest of this article is organized as follows. In Section 2, we present MIMO radar model, and formulate the joint optimization of the WCM and biased estimator. In Section 3, under the weighted or spectral norm constraint on the gradient matrix, we solve the joint optimization problem resorting to the SDP relaxation, and provide a solution to the problem. In Section IV, we assess the effectiveness of the proposed method via some numerical examples. Finally, in Section V, we draw conclusions and outline possible for future research tracks.
Throughout the article, matrices and vectors are denoted by boldface uppercase and lowercase letters, respectively. We use {·}^{T}, {·}^{*}, and {·}^{H}to denote the transpose, conjugate, and conjugate transpose, respectively. vec{·} is the vectorization operator stacking the columns of a matrix on top of each other, I denotes the identity matrix, and ⊗ indicates the Kronecker product. The trace, real, and imaginary parts of a matrix are denoted by tr{·}, Re{·}, and Im{·}, respectively. The symbol {·}^{†} denotes MoorePenrose inverse of a matrix, and {·}^{+} indicates the positive part of a real number. The notation E{·} stands for the expectation operator, diag{a} for a diagonal matrix with its diagonal given by the vector a, and for the Frobenius norm of the matrix A. Given a vector function , we denote by the k × n matrix the ij th element of which is . is the range space of a matrix A. Finally, the notation means that BA is positive semidefinite.
2 System model and problem formulation
Consider a MIMO radar system with M_{t} transmitting elements and M_{r} receiving elements. Let be the transmitted waveform matrix, where ,i = 1,2,...,M_{t} denotes the discretetime baseband signal of the i th transmit element with L being the number of snapshots. Under the assumption that the transmitted signals are narrowband and the propagation is nondispersive, the received signals by MIMO radar can be expressed as
where the columns of are the collected data snapshots, are the complex amplitudes proportional to the RCSs of the targets with K being the number of targets at the considered range bin, and denote the locations of these targets. The parameters and need to be estimated from the received signal Y. The second term in the right hand of (1) indicates the clutter data collected by the receiver, ρ(θ_{ i }) is the reflect coefficient of the clutter patch at θ_{ i }, and N_{C} (N_{C} ≫ M_{t}M_{r} the number of spatial samples of the clutter. The term W denotes the interference plus noise, which is independent of the clutter. Similar to [7], the columns of W can be assumed to be independent and identically distributed circularly symmetric complex Gaussian random vectors with mean zero and an unknown covariance B. a(θ_{ k }) and v(θ_{ k }) denote, respectively, the receiving and transmitting steering vectors for the target located at θ_{ k }, which can be expressed as
where f_{0} represents the carrier frequency, τ_{ m }(θ_{ k }), m = 1,2,...M_{r} is the propagation time from the target located at θ_{ k }to the m th receiving element, and is the propagation time from the n th transmitting element to the target. Also, a_{c} (θ_{ i }) and v_{c} (θ_{ i }) denote the receiving and transmitting steering vectors for the clutter patch at θ_{ i }, respectively.
For notational simplicity, (1) can be rewritten as
where , which represents the clutter transfer function similar to the channel matrix in [2]. According to Chen and Vaidyanathan and Wang and Lu [33, 34], vec(H_{c}) can be considered as an identically distributed complex Gaussian random vector with mean zero and covariance
In fact, can be explicitly expressed as (see, e.g., [35]):
where , , and . Note that is a positive semidefinite Hermitian matrix [33].
We now consider the constrained biased CRB of the unknown target parameters , where , , , and β_{ I }= Im(β). According to Zvika and Eldar Yonina [29], if , the constrained biased CRB can be written as
where
with d(x) denoting the bias for estimating x. U satisfies:
in which is assumed to have full row rank with g(x) being the equality constraint set on x and U is the tangent hyperplane of g(x) [20].
Following [20, 21], some prior information can be available in array signal processing, for example, constant modulus constraint on the transmitted waveform, and the signal subspace constraints in the estimation of the angleofarrival. Here, we assume that the complex amplitude matrix β = diag(β_{1},β_{2},...,β_{ k }) is known as
Remark
In practice, the parameters of one target can be estimated roughly from the received data by many methods (see, e.g., [36] for more details). Therefore, we can obtain the imprecise knowledge of one target by transmitting orthogonal (or uncorrelated) waveforms before waveform optimization. In this article, our main interest is only to improve the accuracy of location estimation by optimizing transmitted waveforms. One can see from Section 3 that the waveform optimization is based on the FIM F that considers the unknown parameters consisting of the location and complex amplitude (see, (11)(16)). Hence, the estimation of complex amplitude matrix β is regarded as prior information for waveform optimization here.
Following (9), we can obtain , where 0_{2K×K}denotes a zero matrix of size 2K × K. Hence, the corresponding null space U can be expressed as
Based on the discussion above, the Fisher information matrix (FIM) F with respect to x is derived in Appendix A and given by
where
The problem of main interest in this study is the joint optimization of the WCM and bias estimator to improve the performance of parameter estimation by minimizing the constrained biased CRB of target locations. It can be seen from (6) that the constrained biased CRB depends on U, D, and F. In practice, it is not obvious how to choose a particular matrix D to minimize the total variance [23]. Even if a bias gradient matrix is given, it may not be suitable because a biased estimator reduces the variance obtained by any unbiased estimator at the cost of increasing the bias. As a sequence, a tradeoff between the variance and bias should be made, i.e., the biased estimator should be optimized [24]. According to Hero and CramerRao [23], optimizing the bias estimator requires its bias gradient belonging to a suitable class. In this article, two constraints on the bias gradient are considered, i.e., the weighted and spectral norm constraints. In Section 3, with each norm constraint, we treat the joint optimization problem under two design criteria, i.e., minimizing the trace and the largest eigenvalue of the constrained biased CRB.
3 Joint optimization
In this section, we demonstrate how the WCM and bias estimator can be jointly optimized by minimizing the constrained biased CRB. First of all, this problem is considered under the weighted norm constraint.
A. Joint Optimization With the Weighted Norm Constraint
Similar to [28], the weighted norm constraint can be expressed as
where M is a nonnegative definite Hermitian weighted matrix, and γ is a constant which satisfies:
First, we consider this problem by minimizing the trace of the constrained biased CRB, which is referred to as the Traceopt criterion [7]. Under the weighted norm constraint (18) and the total transmitted power constraint, the optimization problem can be formulated as
where the second constraint holds because the power transmitted by each transmitting element is more than or equal to zero [6], and P is the total transmitted power.
It can be seen from (6) that J_{CBCRB} is a linear function of F^{1}, and a quadratic one of D. Moreover, F is a nonlinear function of R_{ S }, which can be seen from (11)(14). As a sequence, this problem is a rather complicated nonlinear optimization one, and hence it is difficult to be treated by convex optimization methods [30–32]. In order to solve it, we make a simplifying assumption that R_{ S }⊗ B^{1} spans the same subspace as , i.e.,
the rationality of which is proved under a certain condition in Appendix B. Under this assumption, according to Horn and Johnson [37], the product of R_{ S }⊗ B^{1} and , denoted by R_{ SC }, is positive semidefinite, i.e.,
With (22), the problem in (20) can be solved by SDP relying on the following lemma [38, pp. 472]:
Lemma 1
(Schur's Complement) Let be a Hermitian matrix with C ≻ 0, then Z ≽ 0 if and only if ΔC ≽ 0, where ΔC is the Schur complement of C in Z and is given by ΔC = AB^{H}C^{1}B.
Using Lemma 1, the proposition 1 below can reformulate the nonlinear objective in (20) as a linear one, and give the corresponding linear matrix inequality (LMI) formulations of the first two constraints, which is proved in Appendix C.
Proposition 1
Using matrix manipulations, the first two constraints in (20) can be converted into the following LMIs:
where
and τ, β are given in (75) and (87), respectively. According to Lemma 1, the matrix must be positive definite, which can be guaranteed by (72). From (11)(14) and (25), it is known that the nonlinear objective in (20) can be converted into a linear one with respect to E.
With (6), (23) and (24), the problem (20) can be equivalently represented as
where t is an auxiliary variable.
It is noted that the terms in the left hand of the first two constraint inequalities in (26) are quadratic functions of D, and hence these inequalities are not LMIs. The Proposition 2 below can give the LMI formulations of these inequalities, which is proved in Appendix D.
Proposition 2
Using Lemma 1 and some matrix lemmas, the first two constraint inequalities in (26) can be, respectively, expressed as
Now, the joint optimization problem (20) can be readily cast as an SDP
Next, the joint optimization problem is treated by minimizing the largest eigenvalue of the constrained biased CRB, which is referred to as the Eigenopt criterion [7]. Similar to the case of the Traceopt criterion, the problem can be expressed as
Using Lemma 1 and the results above, this problem is equivalent to SDP as
B Joint Optimization With the Spectral Norm Constraint
The spectral norm constraint, similar to [28], can be written as
where T is a nonnegative definite Hermitian matrix, and γ is a constant satisfying:
with λ_{max} (T) denoting the largest eigenvalue of T.
First, we consider the traceopt criterion. Under the spectral norm constraint (32), the problem can be similarly written as
Following Lemma 1 and the propositions above, (34) can be recast as SDP
Second, similar to the discussion above, the optimization problem under the Eigenopt criterion can be represented as SDP
After obtaining the optimum E from (29), (31), (35), and (36), the term R_{ SB }= R_{ S }⊗ B^{1} can be solved via (25), which can be reshaped as
From (37), we have
Scale R_{ SB } such that
where α is a scalar which satisfies the equality constraint.
Given R_{ SB }, R_{ S }can be constructed via a suitable approximation to it (in a LS sense), which is formulated as
The problem above can be equivalently represented as
Using Lemma 1, (41) can be equivalently represented as an SDP
Using many wellknown algorithms (see, e.g., [30–32]) for solving SDP problems, the problems in (29), (31), (35), (36), and (42) can be solved very efficiently. In the following examples, the optimization toolbox in [32] is used for these problems. It is noted that we only obtain the WCM other than the ultimate transmitted waveforms in this article. In practice, the ultimate waveforms can be asymptotically synthesized by using the method in [39].
4 Numerical examples
In this section, some examples are provided to illustrate the effectiveness of the proposed method as compared with the uncorrelated transmitted waveforms (i.e., R_{ S }= ( P / M_{ t })I).
Consider a MIMO radar system with M_{ t }= 5 transmitting elements and M_{ r }= 5 receiving elements. We use the following two MIMO radar systems with various antenna configurations: MIMO radar (0.5, 0.5), and MIMO radar (2.5, 0.5), where the parameters specifying each radar system are the interelement spacing of the transmitter and receiver (in units of wavelengths), respectively. Let the weighted matrix M = I and γ = 1 in the case of the weighted norm constraint, and T = I and γ = 0.5 in the other case. In the following examples, two targets with unit amplitudes are considered, which are located, respectively, at θ_{1} = 0^{o} and θ_{2} = 13^{o} for MIMO radar (0.5, 0.5), and θ_{1} = 0^{o}and θ_{2} = 7^{o} for MIMO radar (2.5, 0.5). The number of snapshots is L = 256. The array signaltonoise ratio (ASNR) in the following examples varying from 10 to 50 dB is defined as , where denotes the variance of the additive white thermal noise. The clutter is modelled as N_{c} = 10000 discrete patches equally spaced on the range bin of interest. The RCSs of these clutter patches are modelled as independent and identically distributed zero mean Gaussian random variables, which are assumed to be fixed in the coherent processing interval (CPI). The cluttertonoise ratio (CNR) is defined as , which ranges from 10 to 50 dB. There is a strong jammer at 11^{°} with an array interferencetonoise ratio (AINR) equal to 60 dB, defined as the product of the incident interference power and M_{r} divided by . The jammer is modeled as point source which transmits white Gaussian signal uncorrelated with the signals transmitted by MIMO radar.
From Section 3, it is known that the joint optimization problem is based on the CRB that requires the specification of some parameters, e.g., the target location and clutter covariance matrix. In practice, the target parameters and clutter covariance can be estimated by using the method in [36, 35], respectively.
In order to examine the effectiveness of the proposed method, we will focus on the following three cases: the CRB of two angles with exactly known initial parameters, the effect of the optimal biased estimator or prior information on the CRB, and the effect of the initial parameter estimation errors on the CRB.
A.The CRB Without Initial Estimation Errors
Figure 1 shows the optimal transmit beampatterns under the Traceopt criterion in the case of ASNR = 50 dB and CNR = 10 dB. It can be seen that a notch is placed almost at the jammer location. Moreover, the difference between the powers obtained by two targets is large because only the total CRB is minimized here excluding the CRB of every parameter. As a sequence, for a certain parameter, the CRB obtained by the optimal waveforms may be larger than that of uncorrelated waveforms.
Figure 2 shows the CRB of two angles as a function of ASNR or CNR. One can see that the CRB obtained by our method or uncorrelated waveforms decreases as the increasing of ASNR, while increases as the decreasing of CNR. Moreover, the CRB under the Traceopt or Eigenopt criterion is much lower than that of uncorrelated waveforms, regardless of ASNR or CNR. Furthermore, under the same norm constraint, the Traceopt criterion leads to a lower total CRB than the Eigenopt criterion. Besides, by comparing Figure 2a with 2c or Figure 2b with 2d, it follows that the total CRB for MIMO radar (2.5, 0.5) is lower than that for MIMO radar (0.5, 0.5). This is because the virtual receiving array aperture for the former radar is much larger than that for the latter [3].
B.Effect of the Optimal Biased Estimator or Prior Information on the CRB
In this subsection, we will study the CRB obtained by only using the optimal biased estimator or prior information.
First, only the optimal biased estimator is employed. In this case, let the matrix u in (6) be equal to I (All other parameters are the same as the previous examples.). The variant of the CRB for this case is the biased CRB as mentioned above. Figure 3 shows the CRB in this case as a function of ASNR or CNR. It can be seen that the optimal biased estimator may lead to a little higher CRB than using the uncorrelated waveforms sometimes, which is because the total CRB of the amplitudes of two targets is not taken into account here. Moreover, the Traceopt criterion leads to higher improvement of the CRB than the Eigenopt one under the same norm constraint, which is similar to the results obtained from Figure 2.
Second, we examine the CRB obtained by only using the prior information. In this case, let the matrix D in (6) be equal to 0_{3k× 3k}and all the other parameters remain the same as the previous examples. The variant of the CRB for this case is the constrained CRB as stated above. Figure 4 shows the CRB in the case as a function of ASNR or CNR. One can observe that the contributions of the prior information to two optimization criteria are almost identical, and the prior information can significantly improve the accuracy of parameter estimation with the uncorrelated waveforms.
C. Effect of the Initial Parameter Estimation Errors on the CRB
In this subsection, we consider the effect of the initial angle or clutter estimation error on the CRB of two angles. It is noted that the relative error of the clutter estimate is defined as the ratio of the estimation error of the initial total clutter power to the exact one.
Figure 5 shows the CRB versus the estimation error of the initial angle or clutter power with ASNR = 10 dB and CNR = 50 dB under the condition that all the other parameters are exact. We can see that the CRB varies with the estimate error of the angle or clutter very apparently, which indicates that the proposed method is very sensitive to these errors. Hence, the robust method for waveform design is worthy of investigating in the future.
5 Conclusions
In this article, we have proposed a novel constrained biased CRBbased method to optimize the WCM and biased estimator to improve the performance of parameter estimation of point targets in MIMO radar in the presence of clutter. The resultant nonlinear optimization problem can be solved resorting to the SDP relaxation under a simplifying assumption. A solution of the initial problem is provided via approximating to an optimal solution of the SDP one (in a LS sense). Numerical examples show that the proposed method can significantly improve the accuracy of parameter estimation in the case of uncorrelated waveforms. Moreover, under the weighted norm constraint, the Traceopt criterion results in a lower CRB than the Eigenopt one. As illustrated by examples in Section IV, the performance of the proposed method may be degraded when the initial parameter estimates are exploited. One way to overcome this performance degradation is to develop a more robust algorithm for joint optimization against the estimation error, which will be investigated in the future.
Appendix A
Fisher information matrix
Consider the signal model in (3), and stack the columns of Y in a M_{r}L × 1 vector as
Similar to [7], we calculate the FIM with respect to θ, β_{ R }, β_{ I }(Here we only consider onedimensional targets.). According to Xu et al. [40], we have
where Q denotes the covariance of the clutter plus interference and noise, which can be represented as
With (4), (45) can be simplified as
Let h_{k} = v( θ_{ k }) ⊗ a( θ_{ k }). Note that
Because
then
Let . By using matrix inversion lemma, we can get
where R_{ S } = S^{*}S^{T}. With (50), (49) can be rewritten as
and hence
where F_{11} is given in (12).
Similarly, we have
and
Hence
and
where F_{12} is given in (13).
We also have
and
where F_{22} is given in (14).
From (49) and (55)(58), we can obtain (11) immediately.
Appendix B
Proof of the rationality of (21)
It is known that the CRB for an unbiased estimator can be achieved by using the minimum mean square error (MMSE) estimator [27]. Therefore, from the parameter estimation perspective, the optimal transmitted waveforms can be obtained through minimizing the MMSE estimation error. For convenience of derivation, we stack the collected data in (3) into a M_{ r }L × 1 vector as
where h_{t} = vec (H_{ t }), , and h_{c} = vec (H_{c}). In order to minimize the MSE, the optimal MMSE estimator, denoting by G_{op}, should be firstly obtained. According to Eldar Yonina [28], G_{op} can be obtained by solving the following optimization problem:
Differentiating the above function with respect to G and setting it to zero, we have
where . Hence, the MMSE estimate of h_{t} can be represented as:
Accordingly, the MMSE estimation error can be written as
By substituting (61) and (62) into the equation above and using matrix inversion lemma, (63) can be rewritten as
which has the same form as Equation 3 shown in [19]. Therefore, according to Theorem 4 in [19], if and can be joint diagonalized, we can obtain
where Λ_{t} and Λ_{c} are, respectively, the diagonal matrices with each diagonal entry given by a real and nonnegative eigenvalue of and , Q is the unitary eigenvector matrix of and , and μ is a scalar constant that satisfies the transmitted power constraints. It can be seen from (65) that R_{ S }⊗ B^{1} spans indeed the same subspace as . The proof is completed.
Appendix C
Proof of proposition 1
In order to convert the objective in (20) into a linear function, let
then
It is noted that is a Hermitian matrix under the aforementioned assumption. Substituting (66) into (12)(14), we can see that F is the linear function with respect to E. Because
we have
Combining (67) and (69), we can obtain
Hence
Because R_{SC} ≽ 0, we have
From (71), it follows that
Using a wellknown inequality in matrix theory, we have
where
and is the largest eigenvalue of .
With , we can obtain
Using Lemma 1, and (74)(76), (23) follows immediately. In order to obtain (24), we rely on the following lemma.
Lemma 2
Let A and B be positive and nonnegative definite Hermitian matrix, respectively. Then, AB ≽ 0 if AB is a Hermitian matrix.
Proof: According to the similarity property of the matrices [38], AB is similar to a Hermitian matrix . Hence, if we can obtain , then AB ≽ 0. Let y = A^{1/2}x, then
Following the definition of the nonnegative matrix [38], we have
Thus, AB ≽ 0.
Following Lemma 2, it is obvious that
It is noted that R_{ sc } can be diagonalized by its eigenvalue decomposition, i.e.,
where u is a unitary matrix and Σ = diag{λ_{1},λ_{2},...,λMtMr} is a diagonal matrix with each diagonal entry given by a eigenvalue. With R_{ SC }≽ 0, we can obtain λ_{ i }≥ 0, i = 1,2,...,M_{ t }M_{ r }. Then (79) can be rewritten as
Denote the eigenvalue of by . From (81), can be expressed as
From (82), it is known that increases monotonically with λ_{ i }. Hence,
where and λ_{max} are the largest eigenvalues of and R_{ sc }, respectively. As discussion above, we have
Because R_{SC} ≽ 0, then
With (83)(85), we have
where
Then, we have
By combining (79) and (88), (24) follows immediately.
Appendix D
Proof of proposition 2
Because , then
Evidently,
where M^{1/2} is the square root of M [38]. With tr(ABC) = tr(CAB), (90) can be rewritten as
With Lemma 1, (89) and (91), we can obtain (27) and (28).
Abbreviations
 AINR:

array interferencetonoise ratio
 ASNR:

array signaltonoise ratio
 CNR:

cluttertonoise ratio
 CPI:

coherent processing interval
 CRB:

CramerRao bound
 FIM:

Fisher information matrix
 LMI:

linear matrix inequality
 LS:

least squares
 MIMO:

multiinput multioutput
 MMSE:

minimum mean square error
 RCS:

radar cross sections
 SDP:

semidefinite programming
 WCM:

waveform covariance matrix
References
 1.
Fishler E, Haimovich A, Blum R, Chizhik D, Cimini L, Valenzuela R, MIMO radar: an idea whose time has come. In Proceedings of the IEEE Radar Conference. Newark, NJ, USA, 2629; 2004:71.
 2.
Fishler E, Haimovich A, Blum R, Cimini L, Chizhik D, Valenzuela R: Spatial diversity in radarsmodels and detection performance. IEEE Trans Signal Process 2006,54(3):823838.
 3.
J Li, Stoica P: MIMO radar with colocated antennas. IEEE Signal Process Mag 2007,24(5):106114.
 4.
Li J, Stoica P, Xu L, Roberts W: On parameter identifiability of MIMO radar. IEEE Signal Process Lett 2007,14(12):968971.
 5.
Yan H, Li J, Liao G: Multitarget identification and localization using bistatic MIMO radar systems. EURASIP J. Adv Signal Process 2008. Article ID 283483 (2008)
 6.
Peter S, Li J, Xie Y: On probing signal design for MIMO radar. IEEE Trans. Signal Process 2007,55(8):41514161.
 7.
Li J, Xu L, Stoica P, Forsythe KW, Bliss DW: Range compression and waveform optimization for MIMO radar: a CramerRao bound based study. IEEE Trans Signal Process 2007,55(8):41514161.
 8.
Fuhrmann DR, Antonio GS: Transmit beamforming for MIMO radar systems using signal crosscorrelation. IEEE Trans Aerosp Electron Syst 2008,44(1):171186.
 9.
Li J, Xu L, Stoica P, Xiayu Z: Signal synthesis and receiver design for MIMO radar imaging. IEEE Trans Signal Process 2008,56(8):39593968.
 10.
Chen CY, Vaidyanathan PP: MIMO radar ambiguity properties and optimization using frequencyhopping waveforms. IEEE Trans Signal Process 2008,56(12):59265936.
 11.
Liu B, He Z, Zeng J, Liu BY: Polyphase Orthogonal Code Design for MIMO Radar Systems. International Conference on Radar 2006, 1.
 12.
Liu B, He Z, He Q: Optimization of Orthogonal Discrete FrequencyCoding Waveform Based on Modified Genetic Algorithm for MIMO Radar. International Conference on Communication, Circuits, and Systems 2007, 966.
 13.
Friedlander B: Waveform design for MIMO radars. IEEE Trans Aerosp Electron Syst 2007,43(3):12271238.
 14.
Chen CY, Vaidyanathan PP: MIMO radar waveform optimization with prior information of the extended target and clutter. IEEE Trans Signal Process 2009,57(9):35333544.
 15.
Bell MR: Information theory and radar waveform design. IEEE Trans Inf Theory 1993,39(5):15781597. 10.1109/18.259642
 16.
Yang Y, Blum R: MIMO radar waveform design based on mutual information and minimum meansquare error estimation. IEEE Trans Aerosp Electron Syst 2007,43(1):330343.
 17.
Yang Y, Blum R: Minimax robust MIMO radar waveform design. IEEE J Sel Top Signal Process 2007,1(1):147155.
 18.
Leshem A, Naparstek O, Nehorai A: Information theoretic adaptive radar waveform design for multiple extended targets. IEEE J Sel Topics Signal Process 2007,1(1):4255.
 19.
Naghibi T, Namvar M, Behnia F: Optimal and robust waveform design for MIMO radars in the presence of clutter. Signal Process 2010,90(4):11031117. 10.1016/j.sigpro.2009.07.033
 20.
Gorman JD, Hero AO: Lower bounds for parametric estimation with constraints. IEEE Trans Inf Theory 1990,26(6):12851301.
 21.
M , Richard Kozick J, Moore T: Bounds on bearing and symbol estimation with side information. IEEE Trans Signal Process 2001,49(4):822834. 10.1109/78.912927
 22.
Stoica P, Ng Boon C: On the CramerRao bound under parametric constraints. IEEE Signal Process Lett 1998,5(7):177179. 10.1109/97.700921
 23.
Hero AO, CramerRao A: Type Lower Bound for Essentially Unbiased Parameter Estimation. Lincoln Lab., Mass. Inst. Technol., Lexington, MA; 1992. Tech. Rep. 890 DTIC ADA246666
 24.
Hero AO, Fessler JA, Usman M: Exploring estimator biasvariance tradeoffs using the uniform CR bound. IEEE Trans Signal Process 1996, 44: 20262041. 10.1109/78.533723
 25.
Eldar Yonina C, Aharon BT, Arkadi N: Robust meansquared error estimation in the presence of model uncertainties. IEEE Trans Signal Process 2005,53(1):168181.
 26.
Eldar Yonina C, Oppenheim AV: Covariance shaping leastsquares estimation. IEEE Trans Signal Process 2003,51(3):686697. 10.1109/TSP.2002.808125
 27.
Eldar Yonina C: Minimax MSE estimation of deterministic parameters with noise covariance uncertainties. IEEE Trans Signal Process 2006,54(1):138145.
 28.
Eldar Yonina C: Minimum variance in biased estimation: bounds and asymptotically optimal estimators. IEEE Trans Signal Process 2004,52(7):19151930. 10.1109/TSP.2004.828929
 29.
Zvika BH, Eldar Yonina C: On the constrained CramérRao bound with a singular fisher information matrix. IEEE Signal Process Lett 2009,16(6):453456.
 30.
BenTal A, Nemirovski A: Lectures on Modern Convex Optimization, Series on Optimization. MPSSIAM, Philadelphia; 2001.
 31.
Vandenberghe L, Boyd S: Semidefinite programming. SIAM Rev 1996,38(1):4095.
 32.
Lofberg J, YALMIP: A toolbox for modeling and optimization in MATLAB. In Proceedings of the CACSD Conference. Taipei, Taiwan; 2004:284.
 33.
Chen CY, Vaidyanathan PP: MIMO radar spacetime adaptive processing using prolate spheroidal wave functions. IEEE Trans Signal Process 2008,56(2):623635.
 34.
Wang G, Lu Y: Clutter rank of STAP in MIMO radar with waveform diversity. IEEE Trans Signal Process 2010,58(2):938943.
 35.
Ward J: SpaceTime Adaptive Processing for Airborne Radar MIT. Lincoln Lab., Lexington; 1994. Tech. Rep. 1015
 36.
Stoica P, Moses RL: Spectral Analysis of Signals. PrenticeHall, Englewood Cliffs; 2005.
 37.
Horn RA, Johnson CR: Matrix Analysis. Cambridge University Press, Cambridge; 1985.
 38.
ütkepohl HL: Handbook of Matrices. Wiley, New York; 1996.
 39.
Stoica P, Li J, Zhu X: Waveform synthesis for diversitybased transmit beam pattern design. IEEE Trans Signal Process 2008,56(6):25932598.
 40.
Xu L, Li J, Stoica P: Target detection and parameter estimation for MIMO radar systems. IEEE Trans Aerosp Electron Syst 2008,44(3):927939.
Acknowledgements
The authors would like to thank Dr. Magnus Jansson and the anonymous reviewers for their thoughtful and tothepoint comments and suggestions which greatly improved the manuscript. This study is sponsored in part by NSFC under Grant 60825104, Program for Changjiang Scholars and Innovative Research Team in University under Grant IRT0954, and the Major State Basic Research Development Program of China (973 Program) under Grant 2010CB731903, 2011CB707001.
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Wang, H., Liao, G., Liu, H. et al. Joint optimization of MIMO radar waveform and biased estimator with prior information in the presence of clutter. EURASIP J. Adv. Signal Process. 2011, 15 (2011). https://doi.org/10.1186/16876180201115
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Keywords
 Multiinput multioutput (MIMO) radar
 waveform optimization
 clutter
 constrained biased CramerRao bound (CRB)
 Semidefinite programming (SDP)